Simple Class A Amplifier
A 10-W Design giving subjectively better
results than class B transistor amplifiers
by J. L. Linsley Hood, M.I.E.E.
Notes by Rod Elliott
During the past few years a number of excellent
designs have been published for domestic audio amplifiers. However, some
of these designs are now rendered obsolescent by changes in the availability
of components, and others intended to provide levels of power output which
are in excess of the requirements of a normal living room. Also, most designs
have tended to be rather complex.
In the circumstances it seemed worth while
to consider just how simple a design could be made which would give adequate
output power together with a standard of performance which was beyond reproach,
and this study has resulted in the present design.
Output power and
In view of the enormous popularity of the
Mullard "5-10" valve amplifier, it appeared that a 10-watt output would
be adequate for normal use; indeed when two such amplifiers are used as
a stereo pair, the total sound output at full power can be quite astonishing
using reasonably sensitive speakers.
* For today's speakers
and expectations, this is clearly not the case. 10 Watts is likely to be
sufficient for tweeters in a tri-amped system however, and this is the
reason for publication of this circuit.
The original harmonic distortion standards
for audio were laid down by D. T. Williamson in a series of articles published
in Wireless World in 1947 and 1949; and the standard, proposed by him,
for less than 0.1% total harmonic distortion at full rated power output,
has been generally accepted as the target figure for high-quality audio
power amplifiers. Since the main problem in the design of valve audio amplifiers
lies in the difficulty in obtaining adequate performance from the output
transformer, and since modern transistor circuit techniques allow the design
of power amplifiers without output transformers, it seemed feasible to
aim at a somewhat higher standard, 0.05% total harmonic distortion at full
output power over the range 30Hz-20kHz. This also implies that the output
power will be constant over this frequency range.
The first amplifier circuit of which the
author is aware in which a transformerless transistor design was used to
give a standard of performance approaching that of the "Williamson" amplifier,
was that published in Wireless World in 1961 by Tobey and Dinsdale. This
employed a class B output stage, with a series connected transistors in
quasi-complementary symmetry. Subsequent high-quality transistor power
amplifiers have largely tended to follow the design principles outlined
in this article.
The major advantage of amplifiers of this
type is that the normal static power dissipation is very low, and the overall
power-conversion efficiency is high. Unfortunately there are also some
inherent disadvantages due to the intrinsic dissimilarity in the response
of the two halves of the push pull pair (if complementary transistors are
used in asymmetrical circuit arrangement) together with some cross-over
distortion due to the I c /V b characteristics. Much has been done, particularly
by Bailey, to minimise the latter.
An additional characteristic of the class
B output stage is that the current demand of the output transistors increases
with the output signal, and this may reduce the output voltage and worsen
the smoothing of the power supply, unless this is well designed. Also,
because of the increase in current drive with output power, it is possible
for a transient overload to drive the output transistors into a condition
of thermal runaway, particularly with reactive loads, unless suitable protective
circuitry is employed. These requirements have combined to increase the
complexity of the circuit arrangement, and a well designed low- distortion
class B power amplifier is no longer a simple or inexpensive thing to construct.
* The thermal runaway
referred to is now known to be secondary breakdown, where the transistor
suffers from a localised heating on the silicon die. This effect is very
rapid, and can lead to almost instantaneous destruction of a transistor.
This is one reason that MOSFETs are preferred by many amplifiers designers
(the editor is not one of these!).
An alternative approach to the design of a
transistor power amplifier combining good performance with simple construction
is to use the output transistors in a class A configuration. This avoids
the problems of asymmetry in quasi- complementary circuitry, thermal runaway
on transient overload, crossover distortion and signal-dependent variations
in power supply current demand. It is, however less efficient than a class
B circuit, and the output transistors must be mounted on large heat sinks.
The basic class A construction consists
of a single transistor with a suitable collector load. the use of a resistor,
as in Fig 1(a), would be a practical solution, but the best power-conversion
efficiency would be about 12%. An l.F. choke, as shown in Fig1(b), would
give much better efficiency, but a properly designed component would be
bulky and expensive, and remove many of the advantages of a transformerless
design. The use of a second, similar, transistor as a collector load, as
shown in Fig 1(c), would be more convenient in terms of size and cost,
and would allow the load to be driven effectively in push-pull if the inputs
of the two transistors were of suitable magnitude and opposite in phase.
This requirement can be achieved if the driver transistor is connected
as shown in Fig. 2.
This method of connection also meets one
of the most important requirements of a low distortion amplifier :- that
the basic linearity of the amplifier should be good, even in the absence
of feedback. Several factors contribute to this. There is the tendency
of the Ic / Vb non-linearity of the characteristics of the output transistors
to cancel, because during the part of the cycle in which one transistor
is approaching cut-off the other is turned full on. There is a measure
of internal feedback around the loop Tr1 Tr2 Tr3 because of the effect
which the base impedance characteristics of Tr1 have on the output current
of Tr3. Also, the driver transistor Tr3, which has to deliver a large voltage
swing, is operated under conditions which favour low harmonic distortion
:- low output load impedance, high input impedance.
* A potentially worthwhile
improvement to this circuit is the addition of a 0.1 ohm resistor in the
emitter circuit of Tr1. This applies local feedback to the entire gain
stage, providing a significant reduction in distortion. If used, this should
be a 5 Watt wirewound type to handle the current.
A practical power amplifier circuit using
this type of output stage is shown in Fig. 3.
* The upper transistor
(Tr2) is operating as a current source, whose output current is modulated.
This allows the circuit to operate at about half the quiescent current
that would be required if no modulation were applied. The values for R1
and R2 must be selected, based on the gain of Tr3. For a 40 Volt supply,
if Tr3 were to have a gain of 50 at 1A, then ...
The open loop gain of the circuit is approximately
600 with typical transistors. The closed loop gain is determined, at frequencies
high enough for the impedance of C3 to be small in comparison to R4, by
the ratio (R3 +R4)/R4. With the values indicated in Fig. 3, this is 13.
This gives a feedback factor of about 160 milliohms.
(R1 + R2) = 20V / 20mA (base
current) = 1000 ohms.One problem with this approach
is that the current provided by Tr2 will vary with temperature. Readers
wishing to experiment with this circuit should ensure that the current
is checked at normal operating temperature (i.e. HOT). There is no mechanism
in the circuit to prevent thermal runaway, other than the use of a suitably
large heatsink. At some point, the circuit should stabilise the quiescent
current. If it does not (and the current continues to increase), then the
heatsink is too small. To ensure a useful life for the transistors, they
should not operate at greater than 50o C, which in normal conditions
should be quite achievable. Since each transistor operates at (or near)
25 Watts, the heatsink for each transistor should have a thermal capacity
of about 1o C / Watt. A better (i.e. larger) heatsink will do
absolutely no harm, and will ensure freedom from thermal runaway.
There is also a newer version
of this amp, but I have no plans to re-publish at any time.
Since the circuit has unity gain at D.C.,
because of the inclusion of C3 in the feedback loop, the output voltage
Ve, is held at the same potential as the base of Tr4 plus the base emitter
potential of Tr4 and the potential drop along R3 due to the emitter current
of this transistor. Since the output transistor Tr1 will turn on as much
current as is necessary to pull Ve down to this value, The resistor R2,
which together with R1 controls the collector current of Tr2, can be used
to set the static current of the amplifier output stages. It will also
be apparent that Ve can be set to any desired value by small adjustments
to R5 and R6. The optimum performance will be obtained when this is equal
to half the supply voltage. (half a volt or so either way will make only
a small difference to the maximum output power obtainable, and to the other
characteristics of this amplifier, so there is no need for great precision
in setting this.)
* Not mentioned is
the purpose of C1 (in conjunction with R1 and R2). This capacitor provides
"bootstrapping", which attempts to maintain a constant voltage across R2.
If the voltage remains constant across a resistor, it follows that the
current through the resistor must also remain constant. The performance
of this circuit will be severely impaired if the value of C1 is too small
- based upon the lowest frequency of operation, and the parallel value
of R1 and R2. For operation down to 20Hz (assuming R1 + R2 = 1000 ohms),
the capacitor should be at least 220uF.
Silicon planar transistors are used throughout,
and this gives good thermal stability and a low noise level. Also, since
there is no requirement for complementary symmetry, all the power stages
can use n-p-n transistors which offer, in silicon, the best performance
and lowest cost. The overall performance at an output level of 10 watts,
or at any lower level, more than meets the standards laid down by Williamson.
The power output and gain/frequency graphs are shown in Figs. 4 and 5,
and the relationship between output power and total harmonic distortion
is shown in Fig. 6. Since the amplifier is a straight-forward class A circuit,
the distortion decreases linearly with output voltage. (This would not
necessarily be the case in a class B system if any significant amount of
cross- over distortion was present.) The analysis of distortion components
at levels of order of 0.05% is difficult, but it appears that the residual
distortion below the level at which clipping begins is predominantly second
Likewise, the reactance of
C1 must be low with respect to the speaker impedance (preferably less than
1/2 of the speaker impedance at the lowest frequency of interest - 20Hz
is assumed). This works out to be about 2,000uF. A working voltage of not
less than 50V is suggested for all electrolytic capacitors, and for optimum
h.f. performance, a 1uF polyester may be paralleled with each electro.
In my experience this is not needed, but many will disagree, so if you
want it, add it.
output and load impedance
Silicon planar NPN transistors have in
general, excellent high frequency characteristics, and these contribute
to the very good stability of the amplifier with reactive loads. The author
has not yet found a combination of L and C which makes the system unstable,
although the system will readily become oscillatory with an inductive load
if R3 is shunted by a small capacitor to cause roll-off at high frequencies.
The circuit shown in Fig. 3 may be used,
with very little modification to the component values, to drive load impedances
in the range 3-15 ohms. However, the chosen output power is represented
by a different current/voltage relationship in each case, and the current
through the output transistors and the output voltage swing will therefore
be different. The peak-voltage swing and mean output current can be calculated
quite simply from the well-known relationship W=I2R and V=IR,
where the symbols have their customary significance. (it should be remembered,
however, that the calculation of output power is based on RMS values of
current and voltage, that these must be multiplied by 1.41 to obtain peak
values, and that the voltage swing measured is the peak to peak voltage,
which is twice the peak value.)
When these calculations have been made,
the peak-to-peak voltage swing for 10 watts power into a 15-Ohm load is
found to be 34.8 volts. Since the two output transistors bottom at about
0.6 volts each, the power supply must provide a minimum of 36 volts in
order to supply this output. For loads of 8 and 3 ohms, the minimum h.t.
line voltage must be 27V and 17V respectively. The necessary minimum currents
are 0.9, 1.2 and 2.0 amps. Suggested component values for operation with
these load impedances are shown in table 1. C3 and C1 together influence
the voltage and power roll-off at low audio frequency performance is desired
than that shown in figs. 4 and 5.
* I suggest a power
supply voltage of 40 Volts, which will actually allow close to 20 Watts
into an 8 ohm load. Make sure that the heatsinks are sufficient to handle
the additional power dissipation caused by the higher supply voltage. Those
suggested in the construction notes are probably too small for operation
at this supply voltage.
Since the supply voltages and output currents
involved lead to dissipation in the order of 17 watts in each output transistor,
and since it is undesirable (for component longevity) to permit high operating
temperatures, adequate heat sink area must be provided for each transistor.
A pair of separately mounted 125mm by 100mm (5" by 4") finned heatsinks
is suggested. This is, unfortunately, the penalty which must be paid for
class A operation. For supplies above 30V Tr1 and Tr2 should be Mj481s
and Tr3 a 2n1613.
If the output impedance of the preamplifier
is more than a few thousand ohms, the input stage of the amplifier modified
to include a simple f.e.t. source follower circuit shown in fig 8. This
increases the harmonic distortion to about 0.12%, and is therefore (theoretically)
a less attractive solution than a better pre- amplifier.
A high frequency roll-off can be obtained,
if necessary by connecting a small capacitor between the gate of the f.e.t
and the negative (earthy) line.
Some experiments were made to determine
the extent to which the circuit performance was influenced by the type
and current gain of the transistors used. As expected the best performance
was obtained when high-gain transistors were used, and when the output
stage used a matched pair. No adequate substitution is known for the 2N697
/ 2N1613 type used in the driver stage, but examples of this transistor
type from three different manufacturers where used with apparently identical
results. Similarly, the use of alternative types of input transistor produced
no apparent performance change, and the Texas Instuments 2N4058 is fully
interchangeable with the Motorola 2N3906 used in the prototype.
The most noteworthy performance changes
were found in the current gain characteristics of the output transistor
pair, and for the lowest possible distortion with any pair, the voltage
at the point from the loudspeaker is fed should be adjusted so that it
is within 0.25 volt of half the supply line potential.
The transistors used in these experiments
were Motorola MJ480/481, with one exception, in which Texas 2S034 devices
were tried. The main conclusion which can be drawn from this is that the
type of transistor used may not be very important, but that if there are
differences in the current gains of the output transistors, it is necessary
that the device with the higher gain shall be used in the position Tr1.
When the distortion components were found
prior to the onset of waveform clipping, these were almost wholly due to
the presence of second harmonics.
The components necessary for a 10 + 10
watt stereo amplifier pair can be conveniently be assembled on a standard
'Lektrokit' 4" X 4.75" s.r.b.p. pin board, with the four power transistors
mounted on external heat sinks. Except where noted the values of components
do not appear to be particularly critical, and 10% tolerance resistors
can certainly be used without ill effect. The lowest noise levels will
however be obtained with good quality components, and with carbon-film
or metal-oxide resistors.
* Metal film resistors
should be used throughout, as these are superior to carbon film types in
all respects. These are generally only available as 1% or better tolerance,
which will not pose any problems.
A suggested form of power supply unit is
shown in Fig. 9(a). Since the current demand of the amplifier is substantially
constant, a series transistor smoothing circuit can be used in which the
power supply output voltage may be adjusted by choice of the base current
input provided by the emitter follower Tr2 and the potentiometer VR1. With
the values of the reservoir capacitor shown in table 3, the ripple level
will be less than 10mV at the rated output current, provided that the current
gain of the series transistor is greater than 40. For output currents up
to 2.5 amps, the series transistors indicated will be adequate, provided
that they are mounted on heat sinks appropriate to their loading.
However, at the current levels necessary
for operation of the 3-ohm version of the amplifier as a stereo pair, a
single MJ480 will no longer be adequate, and either a more suitable series
transistor must be used, such as the Mullard BDY20, with for example a
2N1711 as Tr2, or with a parallel connected arrangement as shown in Fig.
The total resistance in the rectifier "primary"
circuit, including the transformer secondary winding, must not be less
than 0.25 Ohms. When the power supply, with or without an amplifier, is
to be used with an r.f. amplifier-tuner unit, it may be necessary to add
a 0.25uF (160V) capacitor across the secondary windings of T1 to prevent
transient radiation. The rectifier diodes specified are International Rectifier
potted bridge types.
* Although there was no
mention of this in the original article (and I managed to "lose" the schematic
file for a time), a current limiter was included. This will ensure that
the current through the output devices does not exceed a preset value,
although I believe that the concept is flawed, and is of limited value
in this overall design.
The circuit above shows
the way the current limiter is connected. It will not stabilise the quiescent
(no signal) current, but is only capable of ensuring that the absolute
maximum current does not exceed the value determined by the 100 Ohm pot.
To be useful, a current stabiliser is needed, which will ensure that the
no-signal operating current remains constant regardless of temperature
or supply voltage variations. No information is provided to achieve this
This article (with editorial
notes) is reprinted as a service to readers, who are reminded of copyright
laws, which may restrict the rights of readers for reproduction, commercial
production (etc). The information presented is not intended as a guide
for construction, but is primarily for its interest value, and to serve
as a starting point for other designers.
The original article is now
many years old, and some of the transistor types referred to are now superceded
by vastly better designs. I will leave it to readers to experiment with
device types. While much of the design is still quite relevant to a new
design, I think that this amplifier may be found lacking compared to more
recent design trends. In particular, the biasing system is not stable with
temperature, and DC drift will be evident. In addition, the open-loop gain
is very low, so feedback is far less than might be desirable (although
many will feel that this is a good thing!). As mentioned above, additional
local feedback (0.1 ohm resistor in the emitter of Tr1) may reduce open-loop
distortion, but further reduces the gain. I suggest experimentation (I
have only done some computer simulations so far) and would appreciate feedback
from anyone who tries out this circuit.
I would also suggest that
a single supply power amplifier is not really a proposition for new designs
(although the DoZ uses the same principle), and a bi-polar (+/-) power
supply may be preferable. DC stabilisation then becomes a major issue,
since small DC offset voltages can prove a disaster to tweeters in particular.
The diagrams are not of high
quality, but are the originals from the source WWW page. I do not propose
to redraw these, as this design is provided as information only.
John L Linsley-Hood is a
prolific author of amplifier designs, and still presents new ideas and
circuits in the UK magazine Electronics World (formerly Wireless World).
His influence on the design of quality audio amplifers has been considerable,
and continues to this day. This is not to say that I agree with or endorse
all his ideas or theories, but at least he has the guts to say what he
thinks, and the magazine has the guts to print it, too.